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　　A single-stage power factor correction circuit with constant power control

　　In recent years, power factor correction (pFC) technology has attracted widespread attention. The main disadvantages of traditional two-stage pFC circuits are high cost and complex control circuits. The single-stage power factor correction (SSpFC) converter [1][2][3][4] combines the pFC stage and the DC/DC stage to greatly reduce the cost. However, the SSpFC converter has the problem of excessive DC bus voltage when the load becomes light. Literature [2] uses a feedback coil to reduce the DC bus voltage, but it also reduces the conduction angle of the line current, thereby increasing the total harmonic distortion (THD).

　　In order to solve the above problems and ensure the reduction of DC bus voltage and THD when the load changes, this paper proposes an SSpFC converter with constant power control. The direct energy transfer method allows this circuit to reduce the DC bus voltage without reducing the line current conduction angle. Constant power control allows the output of the converter to be regarded as a voltage source when the output voltage is high, and as a current source when the output voltage is low. When the output voltage changes within a certain range, the output power is approximately constant. 1 Circuit working principle

　　The schematic diagram of a single-stage power factor correction circuit is shown in Figure 1. It is actually a combination of a Boost converter and a flyback converter. The Boost converter operates in DCM mode and can achieve power factor correction when the duty cycle and frequency are constant. The flyback converter can operate in DCM or CCM mode.

　　For the convenience of analysis, it is assumed that the rectified voltage is a constant value in a switching cycle, the capacitor CB is large enough to make the voltage VB basically constant, the flyback transformer is regarded as an ideal transformer, the excitation inductor Lm is connected in parallel on the primary side, and the flyback converter works in CCM mode. The circuit has three working modes as shown in Figure 2, and the main working waveforms are shown in Figure 3. Working mode 1 (t0-t1) At time t0, the switch S is turned on, and the DC bus voltage VB is added to the excitation inductor Lm. Since the flyback converter works in the CCM mode, the linear rise of the current im can be expressed as

　　im=VB/Lm(t-t0)+im(t0)(1)

　　While the inductor Lb works in DCM mode, the current iLb rises linearly from zero, and its expression is:

　　iLb=|Vin|/Lb(t-t0)(2)

　　The current flowing through switch S can be expressed as

　　isw=iLb+im(3)

　　Since the diode Df is reverse biased, no current flows in the coils Ns and Np. Working mode 2 (t1-t2) switch S is turned off at time t1, the diode Df is forward biased, the voltage on the excitation inductor Lm is nVo (where n=Np/Ns), then the linear decrease of the current im can be expressed as

　　im=-nVo/Lm(t-t1)+im(t1)(4)

　　The drain-source voltage VDS on the switch S is VB + nVo, and the current iLb on the inductor Lb flows through the coil Np and the capacitor CB and decreases linearly, and its expression is:

　　iLb=-(VB+nVo-|Vin|)/Lb(t-t1)+iLb(t1)(5)

　　Therefore, the current flowing through the primary coil Np and the secondary coil Ns can be expressed as

　　ip=iLb+im(6)

　　is=nip=n(iLb+im) (7) It can be seen from equation (7) that the secondary current consists of two parts. The load not only obtains energy from the excitation inductor Lm but also directly obtains energy from the inductor Lb, which means that a part Energy can be transferred directly to the load without passing through the energy storage capacitor CB, therefore, the efficiency is greatly improved and the DC bus voltage is reduced.

　　Working mode 3 (t2-t3) The current iLb drops to zero at t2, the diode Db is reverse biased, and the excitation current continues to decrease linearly with a slope nVo/Lm until the switch S is turned on again at t3. At this time, the currents on the primary coil Np and the secondary coil Ns can be expressed as:

　　ip=im(8)

　　is=nip=nim (9) 2 constant power control method

　　Figure 4 shows the block diagram of constant power control. In the figure, KVV and KIIo are the voltage sampling value and current sampling value respectively. Through the voltage division of resistors R3 and R4, the forward input voltage of the first operational amplifier is +, and the signal After amplification, the output voltage of the op amp is obtained. The voltage at this point is equal to the reverse input voltage of the second op amp. According to the virtual short characteristics of the op amp, the output voltage of the first op amp is equal to the voltage of the second op amp. The voltage at the forward input terminal of the op amp is equal, that is, =Vref, from which equation (10) can be obtained.

　　(KiIoR4/R3+R4)+(KVVoR3/R3+R4)=VrefR1/(R1+R2) (10)

　　Assuming a=R2/R1, b=R4/R3, then equation (10) is expressed as

　　(KiI0b/1+b)+(KvV0/1+b)=(Vref)/(1+a) (11)

　　From equation (11), we can get the expression of the output power po as

　　po=VoIo=-(Kv/K1b)Vo2+[Vref(1+b)/K1b(1+a)]Vo (12) From equation (12), we can see that the po~Vo curve is a parabola, at the vertex of the parabola Nearby, the output power po is approximately constant. Taking the output voltage of 80V and the output power of 80W as an example, take KV=0.01, KI=0.1, Vref=5V, so that the vertex of the parabola is located at Vo=80V, po=80W, then a=27.13, b=8.00 can be calculated. So formula (12) can be expressed as

　　po=-0.0125Vo2+2Vo(13)

　　When the output voltage variation range is 60V ~ 100V (±25%), the output power variation is 6.25%.

　　This circuit has the functions of voltage limiting and current limiting at the same time. By transforming equation (11), we can get

　　Io=2-0.0125Vo(14)

　　Vo=160-80Io (15)

　　It can be seen that the current is limited to 2A when the output is short-circuited, and the voltage is limited to 160V when the output is open-circuited.

　　3 Simulation and experimental results

　　Based on the above main circuit and control circuit, simulation and testing were carried out using the following parameters: Lb=300μH, CB=470μF/450V, Lp=Ls=600μH, fs=50kHz, RL=80Ω. Figure 5 shows the input line voltage and line current experimental waveforms; Figure 6 shows the measured circuit efficiency when the input voltage changes. It can be seen that the circuit efficiency can reach more than 82% in a wide input voltage range, which is better than the literature [2][ The efficiency of the circuit proposed in 3] is higher; Figures 7 and 8 show the measurement results of power factor and THD at different input voltages respectively. As can be seen from Figure 7, the power factor of the circuit can be improved when the input voltage is 100~150V. Reaching 0.98, it can also reach 0.96 when the input voltage is 220V; Figure 9 shows the simulation and experimental results of the DC bus voltage VB under different loads when the input voltage is 220V. Both simulation and experiment prove that the DC bus voltage VB changes when the load changes. Can be controlled below 380V.

　　4 Conclusion

　　This paper proposes a single-stage power factor correction circuit with constant power control. The pFC stage of this circuit works in DCM mode with lower THD and higher pF. The circuit's direct energy transfer reduces the DC bus voltage and increases efficiency. The constant power control method makes the circuit have good output characteristics, and the DC bus voltage does not change much when the load changes.

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